Motor driving apparatus and motor system

ABSTRACT

The phase error detection unit PHED detects the phase error PERR between the phase of the BEMF and the phase of the phase switching signal COMM (masking signal MSK) at each of a plurality of detection timings that become the zero crossing timings of the BEMF in the mechanical angular cycle. The PI compensator PICPa has a plurality of cycle setting registers REGN 0_0 to REGN 3_5 for each of a plurality of detection timings, and while switching the registers for each detection timing, the PI compensator determines the cycle setting value NCNTS for bringing the inputted phase error PERR close to zero by reflecting the previous cycle setting value NCNT stored in the register. The clock generation unit CGEN sequentially controls the phase switching signal COMM based on the cycle setting value NCNTS.

CROSS-REFERENCE TO RELATED APPLICATIONS

The disclosure of Japanese Patent Application No. 2018-092980 filed onMay 14, 2018 including the specification, drawings and abstract isincorporated herein by reference in its entirety.

BACKGROUND

The present invention relates to a motor driving apparatus and a motorsystem, and, for example, to a technique for controlling the conductiontiming of a motor.

Japanese unexamined Patent Application publications No. 2003-111485discloses a motor driving control circuit including a current controlloop for controlling a motor current and a PLL control loop fordetermining an conduction timing of the motor. In order to generate theconduction timing synchronized with the back electromotive force (BEMF)by using the PLL control loop, a masking period (non-conduction period)reflecting the conduction timing is provided in a predetermined phase.The PLL control loop detects the zero crossing timing of the backelectromotive force while observing the back electromotive force of apredetermined phase in the masking period, and updates the conductiontiming so that the zero crossing timing is maintained at the center ofthe masking period.

Japanese unexamined Patent Application publications No. 2017-85799discloses a motor driving apparatus for driving a three-phase motor witha sinusoidal wave. The motor driving apparatus divides one cycle of thesinusoidal wave (360 degrees electrical angular) into every 60 deg,fixes the terminal voltage of one phase in the three phases to the powersupply voltage or the ground power supply voltage in each period, andcontrols the terminal voltages of the remaining two phases by the PWMsignal.

SUMMARY

For example, in a motor system such as a hard disk drive (abbreviated asHDD in this specification), it is required to reduce the rotationaljitter of the motor in order to improve the recording density inaccordance with the increase in capacity. In order to reduce therotational jitter, it is necessary to detect the position of the motorwith high accuracy (and thus to generate high-accuracy conduction timingbased on the detected position). For example, as a method of detectingthe position of the sensorless motor, a method of detecting the zerocrossing timing of the back electromotive force within the maskingperiod is known as disclosed in Japanese unexamined Patent Applicationpublications No. 2003-111485.

The position of the motor is normally detected at a plurality ofdetection timings within the mechanical angle of 360° and the motordriving apparatus generates, for example, the conduction timingsynchronized with the plurality of detection timings by PLL (PhaseLocked Loop) control or the like. However, in an actual motor, variousmanufacturing variations such as, for example, variations in theposition of the magnetic pole of the rotor, variations in the mountingposition of the stator or the position sensor, and the like occur. As aresult, variations in the detection timings may occur, which will bereferred to as magnetization variations in the present specification.Since the variation in magnetization causes an error in the conductiontiming via PLL control or the like, it may become a factor forincreasing the rotational jitter.

The embodiments described below have been made in view of the above, andother problems and novel features will be apparent from the descriptionof the present specification and the accompanying drawings.

The motor driving apparatus according to the embodiment includes a phaseerror detecting unit, a compensator, and an conduction timing generatingunit, and controls the targeted rotational phase for control so that thedetected rotational phase obtained by detecting the rotational phase ofthe three-phase motor and the targeted rotational phase for control fordetermining the conduction timing of the three-phase motor aresynchronized. The phase error detection unit detects a phase errorbetween the detected rotation phase and the targeted rotational phasefor control at each of a plurality of detection timings in themechanical angular cycle of the three-phase motor. The compensator has aplurality of registers respectively corresponding to a plurality ofdetection timings, and uses the detected phase error as an input,determines an amount of operation for bringing the phase error close tozero by reflecting the previous state variable, and updates the statevariable. The conduction timing generation unit sequentially controlsthe targeted rotational phase for control based on the determined amountof operation. Here, the compensator stores the state variable in acorresponding one register for each of a plurality of detection timingsin the previous mechanical angular cycle, and determines the amount ofoperation in the current mechanical angular cycle by reflecting thestate variable stored in the corresponding one register for each of aplurality of detection timings in the current mechanical angular cycle.

According to the above-mentioned embodiment, it is possible to reducethe rotational jitter even when the magnetization variation occurs inthe motor.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating a schematic configuration exampleof a motor system according to one embodiment of the present invention.

FIG. 2 is a block diagram illustrating a schematic configuration exampleof a motor driving apparatus according to one embodiment of the presentinvention.

FIG. 3 is a block diagram showing a detailed configuration examplearound the PLL control loop in the block diagram of FIG. 2.

FIG. 4 is a circuit diagram illustrating a detailed configurationexample of a position setting unit in the block diagram of FIG. 3.

FIG. 5 is a timing chart illustrating a schematic example of operationaround the PLL control loop of FIG. 3.

FIG. 6 is a timing chart illustrating an example of operation within oneelectrical angular cycle in FIG. 5.

FIG. 7 is a block diagram illustrating an example of a configuration inwhich the PLL control loop in FIG. 3 is deformed.

FIG. 8 is a block diagram illustrating a schematic configuration exampleof a motor driving apparatus according to the second embodiment of thepresent invention.

FIG. 9 is a circuit diagram illustrating an example of a configurationaround the PLL timing control unit in FIG. 8.

FIG. 10 is a timing chart illustrating an example of the operation ofthe motor driving apparatus of FIG. 8 and FIG. 9.

FIG. 11 is a circuit diagram illustrating a configuration example aroundthe PLL timing control unit in a motor driving apparatus according tothe third embodiment of the present invention.

FIG. 12 is a timing chart illustrating an example of an operation of amotor driving apparatus equipped with a PLL timing control unit in FIG.9.

FIG. 13 is a block diagram illustrating a detailed configuration examplearound the PLL control loop in a motor driving apparatus according to anembodiment 4 of the present invention in FIG. 2.

FIG. 14 is a block diagram illustrating a schematic configurationexample of a motor driving apparatus according to the fifth embodimentof the present invention.

FIG. 15 is a block diagram illustrating a detailed configuration examplearound a PLL control loop in FIG. 14.

FIG. 16 is a timing chart illustrating an example of the operation ofthe motor driving apparatus of FIG. 14 and FIG. 15.

FIG. 17 is a block diagram illustrating a schematic configurationexample of a motor driving apparatus that is a comparative example ofthe present invention.

FIG. 18 is a circuit diagram illustrating an example of a configurationaround a three-phase driver in FIG. 17.

FIG. 19 is a timing chart illustrating an example of operation in anideal state in the motor driving apparatus in FIG. 17.

FIG. 20 is a block diagram illustrating a detailed configuration examplearound a current control loop in FIG. 17.

FIG. 21 is a block diagram showing a detailed configuration examplearound the PLL control loop in FIG. 17.

FIG. 22 is a circuit diagram illustrating a detailed configurationexample of a BEMF detecting unit in FIG. 21.

FIG. 23 is a timing chart illustrating a detailed example of operationin the masking period of FIG. 19.

FIG. 24 is a circuit diagram illustrating a detailed configurationexample of a clock generating unit and a PLL timing control unit in FIG.17.

FIG. 25 is a timing chart illustrating an example of operation whenmagnetization variation occurs in the motor driving apparatus in FIG.17.

DETAILED DESCRIPTION

In the following embodiments, when it is necessary for convenience, thedescription will be made by dividing into a plurality of sections orembodiments, but except for the case where it is specifically specified,they are not independent of each other, and one of them is related tosome or all of modifications, details, supplementary description, andthe like of the other. In the following embodiments, the number ofelements, etc. (including the number of elements, numerical values,quantities, ranges, etc.) is not limited to the specific number, but maybe more than or less than a specific number, except for cases where thenumber is specifically indicated or is clearly limited to the specificnumber in principle.

Furthermore, in the following embodiments, it is needless to say thatthe constituent elements (including element steps and the like) are notnecessarily essential except in the case where they are specificallyspecified and the case where they are considered to be obviouslyessential in principle. Similarly, in the following embodiments, whenreferring to the shapes, positional relationships, and the like ofcomponents and the like, it is assumed that the shapes and the like aresubstantially approximate to or similar to the shapes and the like,except for the case in which they are specifically specified and thecase in which they are considered to be obvious in principle, and thelike. The same applies to the above numerical values and ranges.

The circuit elements constituting the functional blocks of theembodiment are not particularly limited but are formed on asemiconductor substrate such as a single-crystal silicon substrate by anintegrated circuit technique such as a well-known complementary MOStransistor (CMOS).

In all the drawings for explaining the embodiments, the same members aredenoted by the same reference numerals in principle, and repetitivedescriptions thereof are omitted.

First Embodiment

FIG. 1 is a block diagram showing a schematic configuration example of amotor system according to Embodiment 1 of the present invention. FIG. 1shows a configuration example of a hard disk drive (HDD) device as anexample of a motor system. The HDD device of FIG. 1 includes an HDDcontroller HDDCT, a cache memory CMEM, a read/write device RWIC, a motordriving apparatus MDIC, and a disc mechanism DSKM. The HDD controllerHDDCT includes, for example, a system-on-chip (SoC) including aprocessor or the like. The cache memory CMEM and the read/write deviceRWIC are composed of, for example, different semiconductor chips.

The disk mechanism DSKM includes a disk (here, a hard disk) DSK, athree-phase spindle motor (hereinafter, referred to as a three-phasemotor) SPM, a head HD, an arm mechanism AM, a voice coil motor VCM, anda ramp mechanism RMP. The three-phase motor SPM rotationally drives thedisk DSK. The voice coil motor VCM controls the position of the head HDin the radial direction of the disk DSK via the arm mechanism AM. Thehead HD reads and writes data from and to the disk DSK at apredetermined position determined by the voice coil motor VCM. The rampmechanism RMP serves as an evacuation location of the head HD when datareading and writing are not executed.

The motor driving apparatus MDIC is composed of, for example, onesemiconductor chip. The motor driving apparatus MDIC comprises adigital-to-analog converter DAC and a VCM driver VCMDV for driving thevoice coil motor VCM. The motor driving apparatus MDIC includes a PLLcontrol loop LPPLL, a current control loop LPCR, a power controllerOTCT, and a three-phase driver SPMDV for driving the three-phase motorSPM. Further, the motor driving apparatus MDIC includes a serialinterface SIF and a parameter setting register PREG for setting drivingconditions of the three-phase motor SPM and the voice coil motor VCM.

The PLL control loop LPPLL detects the position (rotational phase) ofthe three-phase motor SPM by detecting the back electromotive force ofthe three-phase motor SPM (also referred to as BEMF in thespecification) and generates the conduction timing synchronized with theBEMF by using the PLL control. The current control loop LPCR detects acurrent flowing through the three-phase driver SPMDV by using a sensecurrent generation circuit, sense amplifiers SAs, analog-to-digitalconverters ADCs, and the like, and calculates an error between thedetected current and a current instruction value set in the parametersetting register PREG. The current control loop LPCR determines, foreach PWM cycle, a PWM (Pulse Width Modulation) duty for generatingsinusoidal wave voltages (sinusoidal wave currents) having amplitudesreflecting the calculated error and the sinusoidal wave pattern data tobe incorporated in advance, based on the calculated error and thesinusoidal wave pattern data.

The power control unit OTCT generates a three-phase PWM signal based onthe PWM duty from the current control loop LPCR and uses the three-phasePWM signal to switch and control the three-phase driver SPMDV at anappropriate timing based on the conduction timing from the PLL controlloop LPPLL. As a result, the three-phase motor SPM is driven by athree-phase sinusoidal current synchronized with the BEMF of thethree-phase motor SPM.

The read/write device RWIC drives the head HD and causes the head HD toread and write data. The HDD controller HDDCT controls the entire HDDcontroller. For example, the HDD controller HDDCT communicates with themotor driving apparatus MDIC via the serial interface SIF to instructthe motor driving apparatus MDIC to drive the three-phase motor SPM andthe voice coil motor VCM. The driving condition includes a currentindication for the current control loop LPCR. The HDD controller HDDCTinstructs the read/write device RWIC to read/write data. At this time,write data instructed to the read/write device RWIC and data read fromthe head HD via the read/write device RWIC are held in the cache memoryCMEM.

Next, the overall operation of the HDD device will be briefly described.When the motor driving apparatus MDIC receives the start command of thethree-phase motor SPM from the HDD controller HDDCT, the three-phasemotor SPM (i.e., the disc DSK) is controlled to reach the steady-staterotation at the target rotation speed while increasing the rotationspeed stepwise. In the steady-state rotation, the three-phase motor SPMis driven by a three-phase sinusoidal current. In this case, theamplitudes and phases of the sinusoidal currents are controlled by thecurrent control loop LPCR and the PLL control loop LPPLL, respectively.After the three-phase motor SPM reaches the steady-state rotation, theVCM driver VCMDV moves the head HD onto the disk DSK, and the head HDreads and writes data on the disk DSK.

In such a motor system, in addition to high efficiency, reduction ofrotational jitter is required. In particular, in the HDD device, therecording density can be improved by reducing the rotational jitter whenservo information is written, and thus the capacity can be increased. Inorder to reduce the rotational jitter, it is necessary to detect theposition of the three-phase motor SPM with high accuracy and drive thethree-phase motor SPM at high-precision conduction timing insynchronization with the position.

Therefore, the PLL control loop LPPLL detects the zero crossing timingof the BEMF with high accuracy by using the analog-to-digital converterADC, and generates the conduction timing synchronized with the BEMF bythe PLL control. At this time, the PLL control loop LPPLL is providedwith a position setting unit, which will be described later in detail,in order to generate high-precision conduction timings synchronized withthe BEMF even when magnetization variation occurs. Note that theanalog-to-digital converter ADC in the PLL control loop LPPLL may beshared with the analog-to-digital converter ADC in the current controlloop LPCR to reduce the circuitry size.

Prior to the description of the motor driving apparatus of theembodiment, the configuration and operation of the motor drivingapparatus as a comparative example will be described. FIG. 17 is a blockdiagram showing a schematic configuration example of a motor drivingapparatus as a comparative example of the present invention. FIG. 18 isa circuit diagram showing a configuration example around the three-phasedriver in FIG. 17. FIG. 19 is a timing chart showing an operationexample in an ideal state in the motor driving apparatus of FIG. 17.

As shown in FIG. 1, the motor driving apparatus shown in FIG. 17includes a PLL control loop LPPLL′, a current control loop LPCR, a powercontrol unit OTCT, a three-phase driver SPMDV, a serial interface SIF,and a parameter setting register PREG′. As shown in FIG. 18, thethree-phase driver SPMDV includes a pre-driver unit PDVBK and aninverters unit INVBK. The inverters INVBK include three-phase high-sidetransistors M1 u, M1 v, and M1 w, low-side transistors M2 u, M2 v, andM2 w, and three-phase current detecting circuits IDETu, IDETv, IDETw.The three-phase high-side transistors M1 u, M1 v, and M1 w and thethree-phase low-side transistors M2 u, M2 v, and M2 w are composed of,for example, nMOS transistors or the like.

The three-phase high-side transistors M1 u, M1 v and M1 w are coupledbetween the three-phase driving output terminals OUTu, OUTv, OUTw andthe power supply voltage Vpwr. The three-phase low-side transistors M2u, M2 v and M2 w are coupled between the three-phase driving outputterminals OUTu, OUTv, OUTw and the grounding power supply voltage GND,respectively. Each of the three-phase current detecting circuits IDETu,IDETv, IDETw includes a sensing transistor and the like, detects acurrent flowing through the row-side transistor of each phase, andoutputs a sense voltage Vsens proportional to the detected current tothe shared node.

The pre-driver unit PDVBK includes a pre-driver PDVu, PDVv, PDVw of a uphase, a v phase, and a w phase. The u-phase pre-driver PDVucomplementarily drives the high-side transistor M1 u and the low-sidetransistor M2 u in accordance with the u-phase PWM signals PWMu from theoutput control unit OTCT, specifically, the PWM modulation unit PWMMD.In addition, the u-phase pre-driver PDVu drives both the high-sidetransistor M1 u and the low-side transistor M2 u to the off-state inaccordance with the high-impedance signals HIZu of the u-phase from thePWM modulation unit PWMMD and controls the driving output terminal OUTuof the u-phase to the high-impedance state. The high-impedance signalsHIZu are used, for example, to detect the BEMF of the u-phase.

Similarly, the v-phase pre-driver PDVv appropriately controls thehigh-side transistor M1 v and the low-side transistor M2 v in accordancewith the v-phase PWM signal PWMv and the v-phase high-impedance signalHIZv. The w-phase pre-driver PDVw also appropriately controls thehigh-side transistor M1 w and the low-side transistor M2 w in accordancewith the w-phase PWM signal PWMw and the w-phase high-impedance signalHIZw. The three-phase pre-driver PDVu, PDVv, PDVw generates thethree-phase output detecting signal OUTDETu, OUTDETv, OUTDETw by shapingthe actual signal generated at the three-phase driving output terminalOUTu, OUTv, OUTw in a pulse-like manner.

FIG. 18 shows a schematic configuration of the three-phase motor SPM andthe sense current generation circuit SCRG. The three-phase motor SPMincludes a rotor having an 8-pole structure in this example. In thiscase, four cycles of the electrical angle 360° (also referred to aselectrical angular cycles) are included in the cycle of the mechanicalangular 360° (also referred to as mechanical angular cycle) of thethree-phase motor SPM. In view of this electrical angular cycle, thethree-phase motor SPM can equivalently be represented by three-phaseresistors Ru, Rv, Rw, coils Lu, Lv, Lw, and back electromotive force(BEMF) Vbemf (U, V, W)). For example, taking the u-phase as an example,the resistor Ru, the coil Lu, and the back electromotive force Vbemf (U)are coupled in series between the midpoint tap CT and the driving inputterminal INu of the u-phase. The same applies to the v phase and the wphase, and the resistance, coil and back electromotive force of thecorresponding phase are coupled in series between the midpoint tap CTand the driving input terminals INv, INw of the corresponding phase.

The driving input terminals INu, INv and INw are coupled to the drivingoutputs OUTu, OUTv, OUTw, respectively. In this specification, each ofthe driving input terminals INu, INv and INw and the driving outputterminals OUTu, OUTv, OUTw are also referred to as driving terminals.The voltages of the u-phase driving terminals (OUTu, INu), the v-phasedriving terminals (OUTv, INv) and the w-phase driving terminals (OUTw,INw) are referred to as driving terminal voltages Vu, Vv and Vw,respectively and the voltage of the midpoint tap CT is referred to as amidpoint tapped voltage Vct.

The sense current generation circuit SCRG samples the sense voltage Vsenfrom the three-phase driver SPMDV in accordance with the currentsampling signal ISPL from the PWM modulation unit PWMMD. The sensecurrent generation circuit SCRG generates a sense current Isensproportional to the sampled sense voltage Vsens by using a currentamplifier or the like and causes the sense current to flow through thecurrent detecting resistor RNF.

Returning to FIG. 17, the output controller OTCT includes a PWMmodulation unit PWMMD and a PLL timing control unit PLLTC. As describedwith reference to FIG. 11, the PWM modulation unit PWMMD generatesthree-phase PWM signals PWMu, PWMv, PWMw to the three-phase driverSPMDV. As described with reference to FIG. 11, the PLL timing controlunit PLLTC outputs three-phase high-impedance signals HIZu, HIZv, HIZwto the three-phase driver SPMDV, and outputs a current-sampling signalISPL to the sense current generation circuit SCRG.

In addition to these signals, the PLL timing control unit PLLTC outputsa BEMF sampling signal BSPL, a BEMF polarity signal DIR and a maskingsignal MSK. The BEMF sampling signal BSPL determines timings at whichthe BEMF detection unit BFDET samples and holds the BEMF of the targetphase to be detected. The BEMF polarity signal DIR is a signalindicating whether the transition of the BEMF of the target phase to bedetected is a positive direction or a negative direction. The maskingsignal MSK is a signal that is asserted at a predetermined intervalduring which the zero crossing timing of the BEMF of the target phase tobe detected may exist.

The output control unit OTCT further outputs a phase selection signalSEL and a motor phase signal PH. The phase selection signal SEL is asignal for selecting one phase out of three phases and is used todetermine a target phase to be detected in the BEMF detection unitBFDET. The motor phase signal PH is a clock signal having one cycle ofrotation of the electrical angle 360° of the three-phase motor SPM andis used, for example, when the HDD controller HDDCT of FIG. 1 detectsthe rotation speed of the three-phase motor SPM. Specifically, the HDDcontroller HDDCT detects the rotational speed of the three-phase motorSPM based on, for example, the frequency (or period) of the motor phasesignal PH and the number of poles of the three-phase motor SPM. Then,the HDD controller HDDCT calculates a current instruction value (inother words, a torque value) SPNCR for setting the rotation speed to thetarget rotation speed and sets the current instruction value (torquevalue) in the parameter setting register PREG′.

Prior to a detailed explanation of the PLL control loop LPPLL′ and thecurrent control loop LPCR, a schematic operation of them will beexplained with reference to FIG. 19. FIG. 19 shows an operation examplein a period in which the motor driving apparatus drives the three-phasemotor SPM at a steady-state rotation, assuming an ideal three-phasemotor SPM. The motor driving apparatus drives the three-phase drivingterminals with a sinusoidal wave voltage at conduction timingssynchronized with three-phase back electromotive force (BEMF) Vbemf (U,V, W)) whose phases differ from each other by an electrical angle of120°.

FIG. 19 shows the driving terminal voltage Vu of the u phase in thethree phases. The driving terminal voltage Vu is controlled to be asinusoidal wave voltage having the same phase as the back electromotiveforce (BEMF) Vbemf (U)) of the u-phase. However, strictly speaking, themotor driving apparatus controls not the driving terminal voltage butthe driving terminal current and the back electromotive force (BEMF) soas to have the same phase. Here, the current control loop LPCR has afunction of shaping the driving terminal voltage Vu (driving terminalcurrent) into a sinusoidal wave voltage (sinusoidal wave current), and afunction of determining the amplitude and phase of the sinusoidal wavevoltage (sinusoidal wave current) in accordance with the error betweenthe amplitude of sinusoidal wave current (sinusoidal wave voltage) andthe current instruction value, in turn the target rotational speed.

The current control loop LPCR holds in advance soft patterns (SP1, SP2)and a PWM pattern (PWM) defining the transition of the PWM duty for eachPWM period required to generate a sinusoidal wave voltage (sinusoidalwave current). The soft patterns (SP1, SP2) are used in a period inwhich the change of the sinusoidal wave voltage (sinusoidal wavecurrent) is large, and the PWM pattern (PWM) is used in a period inwhich the change of the sinusoidal wave voltage (sinusoidal wavecurrent) is small. The current control loop LPCR generates a sinusoidalwave voltage (sinusoidal wave current) by using such a pattern, anddetermines the amplitude of the sinusoidal wave voltage (sinusoidal wavecurrent) by performing PI control (P: proportional, I: integral) byinputting the error between the driving current of each phase and thecurrent instruction value detected by the sense current generationcircuit SCRG of FIG. 18 to determine the amplitude of the sinusoidalwave voltage (sinusoidal wave current).

The current control loop LPCR determines an advanced angular phase Gdrvof the sinusoidal wave voltage for controlling the sinusoidal wavecurrent and the back electromotive force BEMF in the same phase. Theadvanced angular phase Gdrv is determined by the driving current valueISPN representing the current amplitude detected by the current errordetection unit CERDET and the cycle setting value NCNT generated by thePLL control loop LPPLL′.

On the other hand, the PLL control loop LPPLL′ detects the phase(detected rotation phase) of the BEMF of each phase, and generates aphase switching signal (targeted rotational phase for control) COMMsynchronized with the detected rotation phase by the PLL control. Thephases (in other words, the conduction timings) of the three-phasedriving terminal voltages (driving terminal currents) are controlledbased on the phase switching signal COMM. In order to detect the phaseof the BEMF of each phase, the PLL control loop LPPLL′ uses a pluralityof masking signals MSK provided at predetermined phase intervals (e.g.,at 60° electrical angle intervals) within a mechanical angular cycle(electrical angular cycle) in synchronization with the phase switchingsignal COMM. The masking signal MSK is asserted to the ‘L’ level in apredetermined phase period called a masking period Tmsk for each cycleof the phase switching signal COMM.

In the masking period Tmsk, the driving terminal of the BEMF targetphase to be detected (e.g., u-phase) is controlled to be in ahigh-impedance state using the high-impedance signal (HIZu) described inFIG. 18. As a result, for example, a back electromotive force (BEMF)Vbemf (U) of the u-phase is observed to the driving terminals (OUTu,INu) of the u-phase in the period of the high impedance state (referredto as the high impedance period) and a driving terminal voltage Vu ofthe u-phase based on the current control loop LPCR is applied to thedriving terminals (LPCR, INu) of the u-phase in the period of the highimpedance state (referred to as the high impedance period).

The PLL control loop LPPLL′ controls the phase of the phase switchingsignal COMM (and thus the phase of the masking period Tmsk) so that thezero crossing timing of the BEMF of the phase to be detected (e.g., theu-phase) is maintained at the reference timing (e.g., the center timing)within the masking period Tmsk. The zero crossing timing of the BEMF isthe timing of the rising zero crossing in which the BEMF crosses theintermediate level of the amplitude (the midpoint tapped voltage Vct) tothe high potential side, or the timing of the falling zero crossing inwhich the voltage crosses to the low potential side. The PLL controlloop LPPLL′ monitors the levels of BEMF, for example, using comparatorsduring high-impedance periods, and assert the zero crossing signalsZCOUT when zero crossing timing is detected.

However, in this instance, the assertion level of the zero crossingsignal ZCOUT changes depending on whether the zero crossing signal risesor falls. Therefore, in this embodiment, the BEMF polarity signal DIRfrom the output control unit OTCT is used, and the zero crossing EORsignal ZCEOR is generated based on the polarity signal DIR and the zerocrossing signal ZCOUT. The BEMF polarity signal DIR is controlled to an“L” level in a period in which the BEMF signal is zero crossing from thehigh potential side to the low potential side and is controlled to an“H” level in a period in which the signal is zero crossing from the lowpotential side to the high potential side. In the masking period Tmsk,the zero crossing EOR signal ZCEOR becomes “H” level until the zerocrossing timing and becomes “L” level after the zero crossing timing,regardless of the polarities of the BEMF.

The PLL control loop LPPLL′ detects the phase error between thereference timing (center timing) and the zero crossing timing in themasking period Tmsk using the error counter. In the masking period Tmsk,the error counter performs a count-up operation during the period inwhich the zero crossing EOR signal ZCEOR is at the “H” level andperforms a count-down operation during the period in which the zerocrossing EOR signal ZCEOR is at the “L” level. As a result, the countedvalue at the end of the masking period Tmsk becomes the final phaseerror PERR. The PLL control loop LPPLL′ performs PI control by inputtingthe phase error PERR to calculate a cycle setting value NCNT, reflectsthe cycle setting value NCNT, and generates a phase switching signalCOMM having a period proportional to the period.

In the ideal steady-state rotation state in the ideal motor as shown inFIG. 19, the phase error PERR maintains zero. In this condition, thephase of the BEMF and the phase of the conduction timings are perfectlysynchronized (coincide), so that rotational jitters do not occur. On theother hand, if the zero crossing timing is earlier than the centertiming of the masking period Tmsk, the phase of the BEMF is advanced,and the final phase error PERR is negative. In this case, since thecycle setting value NCNT is reduced by the PI control, the period of thephase switching signal COMM is shortened (in other words, the phase isadvanced), and the masking period Tmsk is also shifted toward the phaseadvancing side in accordance therewith, the PI control is performed sothat the negative phase error PERR is reduced at the time of detectingthe next phase error.

Conversely, if the zero crossing timing is later than the center timing,the phase error PERR is positive. In this case, since the cycle settingvalue NCNT is increased by the PI control, the period of the phaseswitching signal COMM is lengthened (in other words, the phase isdelayed), and the masking period Tmsk is also shifted toward the phaselag side accordingly, the PI control is performed so that the positivephase error PERR is reduced at the time of detecting the next phaseerror. The motor phase signal PH is generated on the basis of the phaseswitching signal COMM controlled in this manner, and becomes a signalrepresenting a cycle of the electrical angle 360° of the three-phasemotor SPM.

Returning to FIG. 17, the PLL control loop LPPLL′ includes a BEMFdetection unit BFDET, a phase error detection unit PHED, a PLL controlunit (specifically, a PI compensator) PICP′, a clock generation unitCGEN, and a PWM fixing unit PWMFC. The BEMF detection unit BFDET detectsthe zero crossing timing of the BEMF in the target phase to be detectedbased on the phase selection signal SEL.

Specifically, the BEMF detection unit BFDET samples the driving terminalvoltage based on the midpoint tapped voltage Vct at a sampling timingbased on the BEMF sampling signal BSPL for each PWM cycle included inthe masking period Tmsk in a state in which the driving terminal of thetarget phase to be detected is controlled to be in a high-impedancestate as described with reference to FIG. 19. The BEMF detection unitBFDET asserts the zero crossing signal ZCOUT when the sampled BEMFbecomes zero. Note that the sampling timing is asserted in a PWM ONperiod within each PWM period for each PWM period.

The phase error detection unit PHED generally detects a phase error PERRbetween a detected rotational phase obtained by detecting the rotationalphase of the three-phase motor SPM and a targeted rotational phase forcontrol for determining the conduction timings of the three-phase motorSPM. In this case, as described with reference to FIG. 19, the detectedrotational phase is the phase of the BEMF, in particular, the phasedetected based on the assertion timing of the zero crossing signalZCOUT. On the other hand, the targeted rotational phase for control isthe phase of the phase switching signal COMM (in other words, theconduction timing), and more specifically, the phase of the referencetiming (center timing) synchronizing with the phase switching signalwithin the masking period Tmsk.

The PI compensator PICP′ sequentially calculates and outputs the cyclesetting value NCNT (i.e., the amount of operation) for bringing thephase error PERR close to zero while sequentially reflecting the phaseerror PERR by the PI control. In other words, the PI compensator PICP′calculates the cycle setting value NCNT for synchronizing the phase ofthe phase switching signal COMM, which is the targeted rotational phasefor control, with the phase of the BEMF, which is the detectedrotational phase. The clock generation unit (conduction timinggeneration unit) CGEN sequentially controls the phase (and thus theperiod (frequency)) of the phase switching signal COMM based on thecycle setting value NCNT.

More specifically, as shown in FIG. 19, the clock generation unit CGENgenerates the phase switching signal COMM having a cycle proportional tothe cycle setting value NCNT and having a cycle corresponding to 60degrees of the cycle having the electrical angle of 360 degrees. Theclock generation unit CGEN generates an error counter clock ERRCLKserving as an operation clock of the phase error detection unit PHED anda set cycle count value DVCNT. The set cycle count value DVCNT is avalue obtained by counting clock cycles in which the length of one cycleis determined based on the cycle setting value NCNT. The period of thephase switching signal COMM is determined, for example, for 32 clockcycles.

The output control unit OTCT determines the conduction timing of thethree-phase motor SPM based on the phase switching signal COMM andgenerates various signals including the masking signal MSK based on thephase switching signal COMM. As a result, as described with reference toFIG. 19, the PLL control loop LPPLL′ controls the phase or frequency ofthe phase switching signal COMM as a feedback loop so that the zerocrossing timing of the BEMF signal is always maintained at the centertiming within the masking period Tmsk. As a result, the power controlunit OTCT can control the three-phase motor SPM at the conduction timingsynchronized with the phase of the BEMF.

Here, in order to synchronize the conduction timing with the BEMF withhigh accuracy and thus to reduce the rotational jitter, the zerocrossing timing of the BEMF needs to be detected with high accuracy.However, the period during which the BEMF detection unit BFDET candetect the BEMF is limited to the period during which both the PWMsignals of the two conduction phases except for the target phase to bedetected (i.e., the non-conduction phase associated with thehigh-impedance signal) are at the on level. Specifically, it is limitedto a period in which one high-side transistor of the two conductionphases and the other low-side transistor of the two conduction phasesare turned on.

The zero crossing timing of the BEMF does not necessarily exist in thisON level period and may exist in a period in which one conduction phasebecomes an OFF level (in other words, between the ON level period of onePWM period [1] and the ON level period of the next PWM period [2]). Inthis instance, since the zero crossing timing of the BEMF is detected inthe on-level period of the PWM cycle [2], a timing error occurs. Inorder to eliminate such a timing error, the PWM fixing unit PWMFC isprovided in this embodiment. The PWM fixing unit PWMFC, which will bedescribed later in detail with reference to FIG. 21 and the like,forcibly fixes the PWM period, in which the zero crossing timing of theBEMF may exist, to the on-level period.

The current control loop LPCR includes a sense current generationcircuit SCRG, a current detection resistor RNF, a sense amplifier SA, ananalog-to-digital converter ADC, a current error detection unit CERDET,a PI compensator PICC, a driving voltage phase generation unit DVPHG anda sinusoidal wave driving voltage generation unit SINPG. The detailedconfiguration and operation of the current control loop LPCR will bedescribed later with reference to FIG. 20 and will be briefly describedhere.

As described with reference to FIG. 18, the sense current Isensgenerated by the sense current generation circuit SCRG flows through thecurrent detecting resistor RNF. The sense current Isens is a currentproportional to the current flowing in each phase of the three-phasemotor SPM. The sense amplifier SA amplifies the voltage across thecurrent detecting resistor RNF and the analog-to-digital converter ADCconverts the output voltage of the sense amplifier SA into a digitalvalue ADCO.

The current error detection unit CERDET calculates a current error CERRbetween the digital signal ADCO and the current instruction value SPNCRand the PI compensator PICC determines a PWM on-count value for bringingthe current error CERR close to zero. The driving voltage phasegeneration unit DVPHG performs so-called advance angle control andcalculates an advanced angular phase Gdrv for compensating the phasedifference between the driving current phase and the driving voltagephase of the three-phase motor SPM. The sinusoidal wave driving voltagegeneration unit SINPG generates the duty instruction values PWMP, SOFTPfor each PWM cycle based on the PWM on-count value from the PIcompensator PICC, the advanced angular phase Gdrv from the drivingvoltage phase generation unit DVPHG and the phase switching signal COMMfrom the PLL control loop LPPLL′.

The parameter setting register PREG′ holds various parameters (Kp1, Kp2,K1, K2, Kcp, Kci, Krev (U, L)), a PWM cycle setting value PCNT, acurrent instruction value SPNCR, a BEMF threshold value Vthb and a zerocrossing mode signal ZCMD. “Kp 1” and “Kp2” are control gains used inthe PI compensator PICP′ for PLL control and “Kcp” and “Kci” are controlgains used in the PI compensator PICC for current control.

Here, the control cycle of the PLL control is at every zero crossingtiming of the BEMF and the control cycle of the current control is atevery PWM cycle. For this reason, for example, the control gains Kp1 andKp2 are determined so that the control band of the PLL control is aboutseveral 100 Hz and the control gains Kcp and Kci are determined so thatthe control band of the current control is about several kHz to 10 kHz.“K1” and “K2” are parameters reflecting the motor constant and are usedto calculate the advanced angular phase edrv. “KrevU, L” are parametersfor duty compensation and is used in the power control unit OTCT. TheBEMF threshold value Vthb and the zero crossing mode signal ZCMD areused in the PWM fixing unit PWMFC and the BEMF detection unit BFDET.

FIG. 20 is a block diagram showing a detailed configuration examplearound the current control loop in FIG. 17. In FIG. 20, the sensecurrent generation circuit SCRG, the current detecting resistor RNF, thesense amplifier SA and the analog-to-digital converter ADC detect thedriving currents of three phases flowing through the three-phaselow-side transistors (and thus the three-phase motor SPM) as describedwith reference to FIG. 18 and output the detected results as digitalvalues ADCO. The current error detection unit CERDET detects a currenterror between the current instruction value SPNCR and the digital valueADCO from the analog-to-digital converter ADC by using the subtractorSB1.

The PI compensator PICC includes an integrator INT and performs PIcontrol by inputting the current error detected by the current errordetection unit CERDET to calculate the PWM duty PWMD reflecting thecurrent error. The PI compensator PICC calculates the PWM on-count valueby multiplying the PWM duty value PWMD by the PWM cycle setting valuePCNT. At this time, the proportional gain Kcp and the integral gain Kciused in the PI control and the PWM cycle setting value PCNT are held inthe parameter setting register PREG′. The PWM cycle setting value PCNTis a value obtained by converting the time of one cycle of the PWMsignal into a count value and the PWM on-count value is a value obtainedby converting the ON period during one cycle of the PWM signal into acount value.

The sinusoidal wave driving voltage generation unit SINPG receives thePWM on-count value from the PI compensator PICC and the phase switchingsignal COMM from the PLL control loop LPPLL′ and generates dutyindication values PWMP, SOFTP for each PWM cycle. The duty instructionvalues PWMP, SOFTP are instruction values for applying a three-phasesinusoidal wave voltage to the three-phase motor SPM and setting theamplitude of the sinusoidal wave voltage to a value corresponding to thePWM on-count value.

More specifically, the motor driving apparatus divides one cycle of thesinusoidal wave (360° electrical angle) into periods of every 60°electrical angle, fixes one phase in the three-phase driving outputterminals OUTu, OUTv, OUTw to the power supply voltage Vpwr or thegrounding power supply voltage GND in each period and controls theremaining two phases by PWM signals, similarly to the method of Japaneseunexamined Patent Application publications No. 2017-85799. Thesinusoidal wave driving voltage generation unit SINPG determines theduty of the remaining two-phase PWM signals for each PWM cycle, i.e. howto change the duty for each PWM cycle within a period of an electricalangle of 60°.

Specifically, the sinusoidal wave driving voltage generation unit SINPGincludes a PWM pattern generation unit (PPG) and a soft pattern (SP1,SP2) generation unit (SPG). As described with reference to FIG. 19, thePWM pattern generation unit PPG and the soft pattern generation unit SPGinclude in advance a normalization table or the like in which a dutypattern for generating a sinusoidal wave is defined. Each of the PWMpattern generation unit PPG and the soft pattern generation unit SPGperforms weighting based on the PWM on-count value on the value of thenormalization table or the like for each PWM cycle to generate the dutyinstruction values PWMP, SOFTP.

The power controller OTCT includes duty correction units DCPp, DCPs anda PWM modulation unit PWMMD. The duty correction unit DCPp detects anerror in the duty generated between the input and output of thethree-phase driver SPMDV and adds a correction value that cancels theerror to the duty instruction value PWMP to generate a corrected dutyinstruction value PWMR.

Specifically, the duty correction unit DCPp receives the outputdetection signal OUTDET from the three-phase driver SPMDV, detects theactual duty, and determines the correction value based on the differencebetween the actual duty and the duty instruction value PWMP.

Further, when the duty instruction value PWMP is larger than the dutydefined by the duty correction parameters KrevU and L, the dutycorrection unit DCPp determines the correction value based on apredetermined arithmetic expression. That is, when the duty instructionvalue PWMP is large, the on/off state of the transistor becomesinsufficient, and therefore, there is a case where a correction valuedifferent from the correction value when the duty instruction value PWMPis small is required.

The duty correction unit DCPp determines the correction value based onthe arithmetic expression. Similar to the duty correction unit DCPp, theduty correction unit DCPs adds a predetermined correction value to theduty instruction value SOFTP to generate a corrected duty instructionvalue SOFTR.

As shown in FIG. 18, the PWM modulation unit PWMMD controls thethree-phase driver SPMDV using three-phase PWM signals PWMu, PWMv, PWMw.When the three-phase motor SPM is driven by the sinusoidal wave voltage,the PWM modulation unit PWMMD fixes the one-phase (e.g. u-phase)high-side transistor or low-side transistor to the on state every timethe electrical angle is 60° based on the phase switching signal COMM,thereby fixing the one-phase driving output terminal (OUTu) to the powersupply voltage Vpwr or the grounding power supply voltage GND. Then, thePWM modulation unit PWMMD generates the remaining two-phase (v-phase,w-phase) PWM signals (PWMv, PWMw) based on the corrected dutyinstruction values PWMR, SOFTR.

In this manner, the PWM modulation unit PWMMD controls the three-phasedriver SPMDV while performing switching at every electrical angle of60°. Further, since the driving current of the three-phase motor SPM hasa sinusoidal wave shape, the current detected by the current detectingresistor RNF includes a pulsating component in which a cycle of anelectrical angle of 60° including the vertex of the sinusoidal wave isrepeated, as shown in FIG. 20. On the other hand, the currentinstruction value SPNCR is a DC component. Therefore, when the currentinstruction value SPNCR is used as it is, there is a possibility that acurrent error with respect to the digital value ADCO cannot be detectedwith high accuracy. Therefore, the current error detection unit CERDETincludes an instruction current correcting unit CRCP for generatingdigital patterns obtained by copying the sinusoidal wave waveforms and apeak value storing unit PKHD.

The current error detection unit CERDET multiplies the currentinstruction value SPNCR by the digital pattern from the instructioncurrent correcting circuit CRCP using the multiplier MUL1 and outputsthe multiplied current instruction value SPNCR-M to the subtractor SB1.Further, the instructed current correcting circuit CRCP outputs thetrigger signal UPADC at the peak timing of the sinusoidal wave waveformand the peak value storing unit PKHD latches the digital value ADCO asthe driving current value ISPN in accordance with the trigger signalUPADC. The driving current value ISPN represents the amplitude of thedriving current.

The driving voltage phase generation unit DVPHG calculates the advancedangular phase edrv by performing a predetermined calculation using thecycle setting value NCNT from the PLL control loop LPPLL′, the drivingcurrent value ISPN from the current error detection unit CERDET and theparameters K1 and K2 from the parameter setting register PREG′. Thesinusoidal wave driving voltage generation unit SINPG shifts the PWMpattern and the soft pattern by the electrical angle based on theadvanced angular phase edrv and generates the duty instruction valuesPWMP, SOFTP using the shifted pattern.

In other words, in the three-phase motor SPM, the phase of the BEMF andthe phase of the driving current are matched with each other, wherebythe three-phase motor SPM can be driven highly efficient. However, sincethe actual operation is driven by a voltage, the driving voltage phaseneeds to be controlled in order to make the phase of the BEMF coincidewith the phase of the driving current. More specifically, control(referred to as advance angle control) is required to apply the drivingvoltage to the three-phase motor SPM in a phase advanced by the advancedangular phase edrv from the BEMF phase in accordance with variouscoefficients (a resistive component, an inductance component, arotational speed, a back electromotive force constant and a drivingcurrent) of the three-phase motor SPM.

Therefore, the driving voltage phase generation unit DVPHG calculatesthe advanced angular phase edrv based on a prescribed arithmeticexpression using these various coefficients as variables. In this case,the resistive component, the inductance component and the backelectromotive force constant are determined by the parameters K1 and K2,the rotational speed is determined by the cycle setting value NCNT andthe driving current is determined by the driving current value ISPN.

FIG. 21 is a block diagram showing a detailed configuration examplearound the PLL control loop in FIG. 17. FIG. 22 is a circuit diagramshowing a detailed configuration of the BEMF detecting unit shown inFIG. 21. In FIG. 22, the BEMF detection unit BFDET includes adifferential amplifier circuit DAMP, a sample-and-hold circuit SH, anamplifier circuit SAMP, a low-pass filter LPF, a comparator CMPzc and abypass switch SWb.

The differential amplifier DAMP includes an operational amplifier OPA1and a plurality of sets of resistor elements R1 and R2 and amplifies thedriving terminal voltage (Vu or Vv or Vw) of the phases selected by thephase selection signal SEL with a gain based on the resistance values ofthe resistor elements R1 and R2 with reference to the midpoint tappedvoltage Vct.

The sample-and-hold circuit SH samples the output voltage of thedifferential amplifier circuit DAMP according to the BEMF samplingsignal BSPL and holds the output voltage of the differential amplifiercircuit DAMP in the capacitor Csh. The BEMF sampling signal BSPL isasserted every PWM period within the masking period Tmsk. The amplifierSAMP includes an operational amplifier OPA2 and resistive elements R3and R4 and amplifies the voltage held by the capacitive element Csh witha gain based on the resistance values of the resistive elements R3 andR4. The amplifying circuit SAMP is provided to improve the sensingsensitivitie of the subsequent comparator CMPzc. However, in some cases,the amplifier circuit SAMP may be omitted.

The low-pass filter LPF includes a resistive element Rft and acapacitive element Cft and filters the BEMF Vbf amplified by theamplifier SAMP. The low-pass filter LPF is provided to reduce a samplingerror caused by BEMF sampling and smooth the stepwise voltage held bythe capacitive element Csh along with sampling.

The comparator CMPzc compares the output voltage Vo2 of the low-passfilter LPF with the zero voltage and thereby generates a zero crossingsignal ZCOUT. In this embodiment, the comparator CMPzc outputs the “H”level when the output voltage Vo2 is higher than the zero voltage. Also,in this embodiment, the hysteretic comparator is used to preventchattering of the zero crossing signal ZCOUT. The bypass switch SWb iscoupled in parallel to the resistive element Rft of the low-pass filterLPF, bypasses the low-pass filter LPF when it is controlled to be on inaccordance with the zero crossing mode signal ZCMD and does not bypassthe low-pass filter LPF when it is controlled to be off in accordancewith the zero crossing mode signal ZCMD.

In FIG. 21, the phase error detection unit PHED includes EXOR gates EORand error counters ECUNT and generally detects a phase error PERRbetween a detected rotational phase obtained by detecting a rotationalphase of the three-phase motor SPM and a targeted rotational phase forcontrol for determining the conduction timings of the three-phase motorSPM as described with reference to FIG. 17. The EXOR gate EOR performsan EXOR operation between the zero crossing signal ZCOUT from the BEMFdetection unit BFDET and the BEMF polarity signal DIR from the outputcontrol unit OTCT and thereby outputs the zero crossing EOR signal ZCEORas shown in FIG. 19.

The error counter ECUNT operates in the assertion period (maskingperiod) of the masking signal MSK and performs a count-up operation or acount-down operation in accordance with the logical level of the zerocrossing EOR signal ZCEOR using the error counter clock ERRCLK from theclock generation unit CGEN.

For example, as shown in FIG. 19, the error counter ECUNT performs thecount-up operation while the logical level of the zero crossing EORsignal ZCEOR is at the “H” level and performs the count-down operationwhile the logical level of the zero crossing EOR signal ECUNT is at the“L” level. As a result, the phase error PERR obtained as the final countvalue of the error counter ECUNT becomes a negative value when the zerocrossing timing of the BEMF is ahead of the center timing of the maskingperiod Tmsk and becomes a positive value when the zero crossing timingof the PERR is behind the center timing of the masking period ECUNT.

As described with reference to FIG. 17, the PI compensator PICP′sequentially calculates and outputs the cycle setting value NCNT (i.e.the amount of operation) for bringing the phase error PERR close to zerowhile sequentially reflecting the phase error PERR by PI control. Inthis embodiment, the PI compensator PICP′ performs PI control using theprevious phase error PERR held in the phase error register REGP (in theprevious sample period), the previous cycle setting value NCNT held inthe cycle setting register REGN (in the previous sample period) and thecontrol gains Kp1 and Kp2.

At this time, the PI compensator PICP′ reflects the cycle setting valueNCNT in the previous sampling period held in the cycle setting registerREGN to the control gain. As a result, since the control band changes inproportion to the rotational velocity, a wide lock range, optimumresponsiveness and stabilization can be obtained in the PI compensatorPICP′. That is, if the fixed control band is determined in accordancewith the case where the rotation speed is low, the responsiveness in thecase where the rotation speed is high is lowered and conversely, if thefixed control band is determined in accordance with the case where therotation speed is high, the stability in the case where the rotationspeed is low is lowered. The PI compensator PICP′ can be used to solvesuch problems.

The clock generation unit (conduction timing generation unit) CGENsequentially controls the phase (and the frequency) of the phaseswitching signals (targeted rotational phase for control) COMM based onthe cycle setting value NCNT. More specifically, the phase switchingsignal COMM is controlled so that the period becomes longer (thefrequency becomes lower) as the cycle setting value NCNT become larger.The clock generation unit CGEN generates the set cycle count value DVCNTand the error counter clock ERRCLK. The error counter clock ERRCLK is aclock which is adjusted so that the output of the error counter ECUNThas a constant detecting gain with respect to the rotational speed ofthe three-phase motor SPM at all times. Specifically, the error counterclock ERRCLK is controlled so that the cycle becomes longer (thefrequency becomes lower) as the rotational velocity becomes lower (thecycle setting value NCNT becomes larger).

The PWM fixing unit PWMFC includes an analog-to-digital converter ADC, aBEMF level judging unit BLVJG and a PWM fixed sequencer PWMFSQ. Ingeneral, the PWM fixing unit PWMFC first monitors the back electromotiveforce (BEMF) Vbf from the BEMF detection unit BFDET and the zerocrossing signal ZCOUT (in this example, the equivalent zero crossing EORsignal ZCEOR). Then, the PWM fixing unit PWMFC controls the PWM fixedsignal PWMFIX to the asserted level in a period (referred to as anon-fixed period) from a predetermined timing after the amplitude levelof the back electromotive force Vbf becomes smaller than the BEMFthreshold value Vthb until the zero crossing signal ZCOUT (zero crossingEOR signal ZCEOR) is asserted. The PWM modulation unit PWMMD receivesthe PWM fixed signal PWMFIX and fixes the high-side transistor of onephase and the low-side transistor of the other phase of the twoconduction phases to ON in the on-fixed period regardless of the PWMperiod.

Specifically, the analog-to-digital converter ADC converts the backelectromotive force Vbf from the BEMF detection unit BFDET, inparticular, the absolute value of the magnitude of the backelectromotive force Vbf, into a digital value ADCBF.

The BEMF level judging unit BLVJG includes a comparator CMP1 comparesthe digital value ADCBF from the analog-to-digital converter ADC with adigital value corresponding to the BEMF threshold value Vthb and outputsa level detecting signal LVDET as a result of the comparison. Here, whenthe digital value ADCBF (the absolute value of the amplitude of the backelectromotive force Vbf) becomes smaller than the BEMF threshold valueVthb, the comparator CMP1 outputs the level detection signal LVDET of“H” level.

The PWM fixed sequencer PWMFSQ includes AND gates AD1 to AD3, an OR gateOR1 and flip-flops FF1 to FF3. The PWM fixed sequencer PWMFSQ receivesthe level detecting signal LVDET from the comparator CMP1 and the zerocrossing signal ZCOUT (specifically, the zero crossing EOR signal ZCEOR)and generates a PWM fixed signal PWMFIX which becomes an asserted level(here, “H” level) during the on-fixed period.

FIG. 23 is a timing chart showing a detailed operation example in themasking period of FIG. 19. As shown in FIG. 23, the driving terminalvoltage Vu of the non-conduction phase (in this case, u phase) servingas the target phase to be detected changes in synchronization with thePWM signal of the conduction phases (v phase, w phase) of the remainingtwo phases. In a period (referred to as a PWM ON period Ton) in whichthe PWM signal is at the “H” level (ON level), one of the drivingterminal voltages Vv and Vw of the two conduction phases (v-phase andw-phase) becomes the power supply voltage Vpwr when the high-sidetransistor is turned on and the other of the driving terminal voltagesVv and Vw becomes the ground power supply voltage GND when the low-sidetransistor is turned on. As a result, the midpoint tapped voltage Vctbecomes “Vpwr/2” and the driving terminal voltage Vu in thenon-conduction phase becomes a voltage obtained by superimposing theback electromotive force Vbemf (U) on the midpoint tapped voltage Vct.

On the other hand, in a period (referred to as a PWM off period Toff) inwhich the PWM signal is at the “L” level (off level), the refluxoperation is performed on the high side or the low side of the twoconduction phases (v phase, w phase).

FIG. 23 illustrates a case where the reflux operation is performed onthe high side. According to the feedback operation, the driving terminalvoltages Vv and Vw of the two conduction phases become near the powersupply voltage Vpwr when the high-side transistor is turned on, orbecome near the ground power supply voltage GND when the low-sidetransistor is turned on. Therefore, it is not easy to detect the backelectromotive force Vbemf (U) from the driving terminal voltage Vu inthe PWM-off period Toff. Therefore, the zero crossing timing of the BEMFis detected in the PWM on-period Ton within each PWM cycle Tpwm for eachPWM cycle Tpwm.

The BEMF sampling signal BSPL for detecting the BEMF is set to a timingat which ringing is avoided and is set to a timing after the center ofthe PWM cycle Tpwm in this embodiment. In addition, the current samplingsignal ISPL for current control and the operation period of theanalog-to-digital converters ADC associated therewith are also set fromthe center timings of the PWM cycle Tpwm. As a result, the mean value ofthe driving current in the PWM cycle Tpwm can be sampled.

The sample-and-hold circuit SH shown in FIG. 22 samples the drivingterminal voltage Vu of the non-conduction phase with reference to themidpoint tapped voltage Vct in accordance with the assertion level (“H”level) of the BEMF sampling signal BSPL. As a result, the input voltage(Vbf) to the low-pass filter LPF changes in the same manner as the backelectromotive force Vbemf (U) in the “H” level period of the BEMFsampling signal BSPL and becomes a constant level in the “L” levelperiod so that the input voltage Vbf becomes a stepped waveform. Thelow-pass filter LPF smooths an input voltage Vbf having a steppedwaveform to generate an output voltage Vo2.

However, in the method of sampling the BEMF in the PWM ON period Tonwithin the PWM cycle Tpwm including the PWM ON period Ton and the PWMOFF period Toff as described above, an error may occur when, forexample, the zero crossing timing of the BEMF is located in the PWM OFFperiod Toff as described with reference to FIG. 17. Therefore, in thiscase, first, the BEMF level judging unit BLVJG in the PWM fixing unitPWMFC performs determination on the staircase-shaped back electromotiveforce Vbf using the BEMF threshold value Vthb.

The BEMF threshold value Vthb is set to, for example, “2×Dbemf” or thelike, with the change in BEMF occurring in one PWM cycle Tpwm as“Dbemf”. The BEMF threshold value Vthb is obtained in advance, forexample, by simulations or the like, and is set in the parameter settingregister PREG′ from the outside of the device. When the amplitude levelof the back electromotive force Vbf becomes smaller than the BEMFthreshold value Vthb at the end of the converting period (B) by theanalog-to-digital converter ADC, the BEMF level judging unit BLVJGoutputs the level detecting signal LVDET of the “H” level.

When the level detecting signal LVDET becomes the “H” level, thePWM-fixed signal PWMFIX is controlled to the “H” level at the risingedge of the subsequent BEMF sampling signal BSPL. In the period(on-fixed period Tofx) in which the PWM fixed signal PWMFIX is at the“H” level, the PWM signal in the conduction phase is fixed at the “H”level and the BEMF sampling signal BSPL is also fixed at the “H” level.As a result, the BEMF detection unit BFDET of FIG. 22 continuouslydetects the BEMF instead of every PWM cycle in the on-fixed period Tofx.

Accordingly, the back electromotive force Vbf also has a continuouswaveform instead of a stepped waveform.

Thereafter, when the zero crossing signal ZCOUT (the zero crossing EORsignal ZCEOR) is asserted, the PWM-fixed signal PWMFIX is controlled to“L” level. In response to this, the “H” levels of the PWM signal of theconduction phase and the BEMF sampling signal BSPL are also released andthe operation returns to the normal operation. That is, the PWM signalof the conduction phase becomes a signal based on a predetermined dutyinstruction and the BEMF sampling signal BSPL becomes a signal assertedin the PWM on-period Ton every PWM cycle Tpwm.

Such an operation of the PWM fixing unit PWMFC makes it possible todetect the zero crossing timing of the BEMF with high accuracy andconsequently to reduce the rotational jitter.

The bypass switch SWb shown in FIG. 22 bypasses the low-pass filter LPFbased on the zero crossing mode signal ZCMD of the parameter settingregister PREG′ when the zero crossing detecting operation using the PWMfixing unit PWMFC is performed.

This eliminates the delay time associated with the filtering and enablesthe zero crossing timing to be detected with higher accuracy.

On the other hand, since the amplitude of the BEMF changes depending onthe rotation speed of the three-phase motor SPM, the BEMF thresholdvalue Vthb also needs to be changed depending on the rotation speed.Therefore, as shown in FIG. 23, the zero crossing detecting operationusing the PWM fixing unit PWMFC is enabled, for example, at the time ofsteady-state rotation in which rotational jitter is a concern and isdisabled, for example, at the time of starting the three-phase motorSPM. In the disabled state, the zero crossing detection is performed bya method of sampling the BEMF in the PWM ON period Ton in the PWM cycleTpwm including the PWM ON period Ton and the PWM OFF period Toff (amethod of monitoring the step-like back electromotive force Vbf throughthe low-pass filter LPF) as usual. The switching of theenabling/disabling is performed by the zero crossing mode signal ZCMD.

FIG. 24 is a circuit diagram showing a detailed configuration example ofthe clock generation unit and the PLL timing control unit in FIG. 17.The clock generation unit CGEN includes counters CUNTn, CUNTd,comparators CMPn,CMPc1,CMPc2 and an error clock generation unit ERCG.The counter CUNTn counts up by using the reference clock CKref fordigital control as a trigger and the comparator CMPn outputs a pulsesignal when the count value (TCNT) of the counter CUNTn reaches thecycle setting value (NCNT). The count value (TCNT) of the counter CUNTnis reset in response to the pulse signal (reset signal TCNTRST).

The counter CUNTd counts up with the pulse signal outputted from thecomparator CMPn as a trigger and outputs the count value as a set cyclecount value DVCNT. The comparator CMPc2 outputs a pulse signal when theset cycle count value DVCNT reaches a predetermined value, e.g. 32. Theset cycle count value DVCNT of the counter CUNTd is reset in response tothe pulse signal DCNTRST. As a result, the counter CUNTd repeats theoperation of counting 32 cycles of the cycle based on the cycle settingvalue NCNT (sequentially outputting 0 to 31).

The comparator CMPc1 outputs the “H” level during the period in whichthe set cycle count value DVCNT are 0 to 15 and outputs the “L” levelduring the period in which the set cycle count value DVCNT are 16 to 31.The output signal becomes a phase switching signal COMM. The error clockgeneration unit ERCG generates an error counter clock ERRCLK having afrequency corresponding to the cycle setting value NCNT by using thereference clock CKref.

The PLL timing control unit PLLTC includes an adder ADDh, comparatorsCMPh, CMPm1, CMPm2, an AND gate ADh, a NAND gate NDm and apolarity-signal generator DIRG. The comparator CMPm1 outputs the “H”level in the period in which the set cycle count value DVCNT is 24 to 31and the comparator CMPm2 outputs the “H” level in the period in whichthe set cycle count value DVCNT is 0 to 31. The NAND gate NDm assertsthe masking signal MSK to the “L” level in a period in which the setcycle count value DVCNT is 24 to 31 by performing a logical operation onthe outputs of the comparators CMPm1, CMPm2.

The adder ADDh adds “−1” to the determination value (here, 24) of thecomparator CMPm1 and sets the result (i.e. 23) as the determinationvalue of the comparator CMPh. As a result, the comparator CMPh outputsthe “H” level in the period in which the set cycle count value DVCNT is23 to 31. The AND gate ADh asserts the high-impedance signal HIZ to the“H” level in a period in which the set cycle count value DVCNT includingthe masking period is 23 to 31 by performing a logical operation on theoutputs of the comparators CMPh, CMPm2. The polarity signal generationunit DIRG generates the BEMF zero crossing polarity signal DIR byperforming polarity inversion for each rising edge of the phaseswitching signal COMM.

Here, the assertion timing of the high impedance signal HIZ is one countbefore the assertion timing of the masking signal MSK. This is toprevent erroneous detection of the zero crossing timing of the BEMF.That is, when the phase to be detected by the BEMF becomes ahigh-impedance state in accordance with the assertion of thehigh-impedance signal HIZ, there is a possibility that the BEMF zerocrossing occurs due to the kickback. This is to prevent erroneousdetection accompanying this.

FIG. 25 is a timing chart showing an operation example in the case wheremagnetization variation occurs in the motor driving apparatus of FIG.17. In FIG. 25, unlike the case of FIG. 19, magnetization variationoccurs. If magnetization variation occurs in the three-phase motor SPM,the phase error PERR may fluctuate without maintaining zero even whenthe PLL control is locked. For example, in FIG. 25, the magnetization ofthe BEMF of the v phase and the w phase is relatively advanced to be inphase.

In this instance, for example, at the time of detecting the falling zerocrossing of the w phase, the phase error PERR becomes negative and thecycle setting value NCNT becomes small, thereby shortening the cycle ofthe subsequent phase switching signal COMM. At the time of detecting therising zero crossing of the subsequent v phase, since it is also theadvanced phase, the phase error PERR becomes zero with the period of thephase switching signal COMM which becomes shorter and the cycle settingvalue NCNT does not change greatly. At the time of detecting thetrailing crossing of the subsequent u-phase, since the magnetizationvariation does not occur, the phase error PERR becomes positive with theperiod of the shortened phase switching signal COMM and the cyclesetting value NCNT become large. Here, for convenience of explanation,the cycle setting value NCNT immediately follows the phase error PERR,but actually, the cycle setting value NCNT smoothly follows the phaseerror PERR through a certain delay according to the control band of thePI-compensator PICP′.

At the time of the steady-state rotation, if the magnetization variationdoes not occur in the ideal state as shown in FIG. 19, the phase errorPERR inputted to the PI compensator PICP′ becomes zero. In this idealstate, the three-phase motor SPM can be driven in a state in which theBEMF and the conduction timings are always synchronized with each otherso that rotational jitter does not occur. On the other hand, when themagnetization variation occurs as shown in FIG. 25, the phase of theBEMF varies for each cycle of the phase switching signal COMM andaccordingly the phase error PERR also varies as appropriate. This stateis a state in which the synchronization between the BEMF and theconduction timings varies and therefore a rotational jitter is incurred.More specifically, the PI compensator PICP′ operates so as to suppressthe variation of the phase error PERR due to the magnetizationvariation, but in practice, the PI compensator PERR′ can suppress thevariation only to a certain extent. The variation component which cannotbe suppressed becomes a cause of the rotational jitter.

In the case of the position-sensorless three-phase motor SPM as shown inFIG. 17, the magnetization variation is caused by, for example,variation in the position of the magnetic pole of the rotor, variationin the mounting position of the stator or the like. Further, in the caseof a motor with a sensor, it is also caused by variations in themounting position of the position sensor or the like. On the other hand,it is also possible to suppress the occurrence itself of suchmagnetization variation, that is, manufacturing variation and therebyreduce the rotational jitter, but in this case, the manufacturing costis increased. For this reason, any mechanism capable of sufficientlyreducing the rotational jitter is required even when the magnetizationvariation occurs.

FIG. 2 is a block diagram showing an example of a schematicconfiguration of a motor driving apparatus according to Embodiment 1 ofthe present invention. The motor driving apparatus shown in FIG. 2differs from the motor driving apparatus shown in FIG. 17 in theconfiguration of the PLL control unit (PI compensator) PICP in the PLLcontrol loop LPPLL. Along with this, in FIG. 2, a position setting unitPSCT is added.

Further, the parameter setting register PREGa outputs a detection modeselection signal CNTMD and a number of poles setting value NPOLE inaddition to the various signals described with reference to FIG. 17.

The PI compensator PICP has a plurality of cycle setting registerscorresponding to a plurality of detection timings (e.g. BEMF zerocrossing timings) provided within one mechanical angular cycle as willbe described in detail later. The PI compensator PICP stores thedetermined cycle setting value (amount of operation) NCNTS in thecorresponding one cycle setting register for each of a plurality ofdetection timings in the previous mechanical angular cycle. Then, the PIcompensator PICP determines the cycle setting value NCNTS in the currentmechanical angular cycle by reflecting the cycle setting value NCNTSstored in the corresponding one cycle setting register for each of aplurality of detection timings in the current mechanical angular cycle.

The position setting unit PSCT controls switching of the cycle settingregister for each of the plurality of detection timings based on thephase switching signal COMM from the clock generation unit CGEN and thenumber of poles setting value NPOLE from the parameter setting registerPREGa. Specifically, the position setting unit PSCT generates a poleposition signal POLE representing the position of the electrical angle360° (electrical angular cycle) within the mechanical angular cycle anda phase position signal CNT6 representing the position of the section ofthe electrical angle 60° (phase switching section based on the phaseswitching signal COMM) within the electrical angular cycle.

In addition, the position setting unit PSCT generates a write enablesignal WEN for enabling the writing of the corresponding cycle settingregister in accordance with the pole position and the phase position.The PI compensator PICP can select two operation modes (switching modeand fixed mode) according to the detection mode selection signal CNTMD.In the switching mode, the PI compensator PICP calculates the cyclesetting value NCNTS while switching the cycle setting registercorresponding to each of the plurality of detection timings. On theother hand, in the fixed mode, the PI compensator PICP stores the cyclesetting value NCNTS for each of the plurality of detection timingscommonly in the plurality of cycle setting registers, therebycalculating the cycle setting value NCNTS by regarding the plurality ofcycle setting registers as one cycle setting register. Thus, in thefixed-mode, the PI compensator PICP performs the same operation as thePI compensator PICP′ shown in FIGS. 17 and 19.

FIG. 3 is a block diagram showing a detailed configuration examplearound the PLL control loop in FIG. 2. FIG. 4 is a circuit diagramshowing a detailed configuration example of the position setting unit inFIG. 3. The configuration example shown in FIG. 3 differs from theconfiguration example shown in FIG. 21 in the inner configuration of thePI-compensator PICP. Along with this, a position setting unit PSCT isadded. The PI-compensator PICP corresponds to a digital loop filter usedin digital PLL control.

Digital loop filters are known in various configurations and may havevarious state variables therein but include at least a component thatreflects the previous state variable to determine the current amount ofoperation (cycle setting value NCNTS) and updates the state variable toreflect the next amount of operation.

The PI compensator PICPa of FIG. 3 includes a cycle setting registerunit REGNUa for storing a cycle setting value NCNT(NCNTS) as one of thestate variables at every detection timing instead of one cycle settingregister REGN shown in FIG. 21. In this embodiment, it is assumed thatthe 8-pole motor shown in FIG. 18 is used, four electrical angularcycles are included in the mechanical angular cycle and six detectiontimings (BEMF zero crossing timings) are included in each electricalangular cycle for every electrical angle of 60° (phase switchinginterval).

In this instance, the cycle setting register unit REGNUa comprises 24cycle setting registers REGN0_0, 0_1, 0_2, 0_3, 0_4, 0_5, 1_0, . . .3_4, 3_5 corresponding to a total of 24 detection timings within themechanical angular cycle, respectively. “x” in the cycle settingregister REGNx-y represents a pole position (position of the electricalangular cycle) within the mechanical angular cycle and “y” represents aphase position within the electrical angular cycle. The cycle settingregister unit REGNUa includes a selector SELN. The selector SELN selectsa corresponding one of the cycle setting registers from the plurality ofcycle setting registers REGN0_0 to REGN3_5 at each detection timing andoutputs the cycle setting value NCNT stored in the selected cyclesetting register to the clock generation unit CGEN as the cycle settingvalue NCNTS.

For example, it is assumed that the cycle setting register REGN0_0 isselected in accordance with a certain detection timing (t[0]) within themechanical angular cycle (T[0]). In this instance, the cycle settingvalue NCNT stored in the cycle setting register REGN0_0 is outputted asthe cycle setting value NCNTS and the cycle setting value NCNT isupdated via the multiplier and the adder to reflect the cycle settingvalue NCNTS and is written into the cycle setting register REGN0_0.

The cycle setting value NCNT written in the cycle setting registerREGN0_0 are used at the same detection timing (t[0]) within the nextmechanical angular cycle (T[1]). In this manner, the current cyclesetting value NCNTS at a certain detection timing is determined byreflecting the previous cycle setting value NCNT at the same detectiontiming.

The position setting unit PSCT comprises a pole position counter CUNTp1,a phase position counter CUNTph and a position decoder PSDEC as shown inFIG. 3 and in particular, comparators CMPph, CMPpl as shown in FIG. 4.The phase position counter CUNTph counts the period of the phaseswitching signal COMM and outputs the count value as the phase positionsignal CNT6. The comparator CMPph outputs a pulsed signal when the phaseposition signal CNT6 reaches the determination value (here, “6”). Thecount value (CNT6) of the phase position counter CUNTph is reset inresponse to the pulse signal (reset signal CNT6RST).

The pole position counter CUNTp1 counts up by using the pulse signaloutputted from the comparator CMPph as a trigger and outputs the countvalue as a pole position signal POLE. The comparator CMPpl outputs apulse signal when the pole position signal POLE reaches a predetermineddetermination value. The predetermined determination value is a halfvalue of the number of poles setting value NPOLE and is “4” for an8-pole motor.

The count value (POLE) of the pole position counter CUNTp1 is reset inresponse to the pulse signal (reset signal POLERST). As a result, thepole position signal POLE cyclically outputs 0 to 3 and the phaseposition signal CNT6 sequentially outputs 0 to 5 for each of the valuesof the pole position signal POLE.

When selecting the switching mode based on the detection mode selectionsignal CNTMD, the position decoder PSDEC asserts any one of a pluralityof write enable signals WEN (here, 24 write enable signals WEN) based oncombinations of the pole position signal POLE and the phase positionsignal CNT6. On the other hand, when the position decoder PSDEC selectsthe fixed mode based on the detection mode selection signal CNTMD, forexample, the position decoder fixes the pole position signal POLE andthe phase position signal CNT6 to zero and commonly asserts a pluralityof write enable signals WEN.

A plurality (24) of cycle setting registers REGN0_0 to REGN3_5 in FIG. 3respectively store inputted cycle setting value NCNT in response toassertions of a plurality of write enable signals WEN. The selector SELNselect one of the outputs of the cycle setting registers REGN0_0 toREGN3_5 based on combinations of the pole position signal POLE and thephase position signal CNT6.

FIG. 5 is a timing chart showing a schematic operation example aroundthe PLL control loop of FIG. 3. FIG. 6 is a timing chart showing anoperation example within one electrical angular cycle in FIG. 5. Asshown in FIG. 5, in the 8-pole motor, four electrical angular cycles T0to T3 are included in the mechanical angular cycle Tcyc1. The electricalangular cycle T0 includes six phase switching intervals T0_0 to T0_5based on the period of the phase switching signal COMM. In the phaseswitching periods T0_0 to T0_5, the PI control operation using thecorresponding cycle setting registers REGN0_0 to REGN0_5 is performed.Similarly, in the phase switching sections T3_0 to T3_5 (T 3_0 to T3_3are not shown) included in the electrical angular cycle T3, the PIcontrol operation using the corresponding cycle setting registers REGN3_0 to REGN 3_5 is performed.

Here, when the method of the first embodiment is used, conceptually, forexample, the cycle setting registers REGN0_0 to REGN0_5 store the cyclesetting value NCNT reflecting the lengths of the phase switchingsections T0_0 to T0_5, respectively. Then, in the mechanical angularcycle Tcyc2 following the mechanical angular cycle Tcyc1, the length ofthe phase switching section T0_0 included in the electrical angularcycle T0 (cycle of the phase switching signal COMM) is determined byreflecting the cycle setting value NCNT stored in the cycle settingregister REGN0_0 within the mechanical angular cycle Tcyc1.

When the magnetization variation as shown in FIG. 25 occurs, forexample, the interval of the zero crossing timing corresponding to thephase switching section T0_0 in the mechanical angular cycle Tcyc2 maybe different from the interval of the zero crossing timing correspondingto the phase switching section T3_5 in the mechanical angular cycleTcyc1. The PI compensator PICP′ shown in FIG. 21 conceptually determinesthe length of the phase switching interval T0_0 by reflecting theinterval of zero crossing timing corresponding to the phase switchinginterval T3_5. However, since the predetermined length differs from theinterval of the zero crossing timing corresponding to the phaseswitching section T0_0 due to the magnetization variation, a phase errorPERR occurs as shown in FIG. 25.

On the other hand, even when such variation in magnetization occurs, theinterval of the zero crossing timing corresponding to the phaseswitching section T0_0 in the mechanical angular cycle Tcyc2 and theinterval of the zero crossing timing corresponding to the phaseswitching section T0_0 in the mechanical angular cycle Tcyc1 are thesame. The PI compensator PICP shown in FIG. 3 conceptually determinesthe length of the phase switching interval T0_0 in the mechanicalangular cycle Tcyc2 by reflecting the interval of the zero crossingtiming corresponding to the phase switching interval T0_0 in themechanical angular cycle Tcyc1. As a result, the phase error PERR can besuppressed and ideally, can be zeroed.

FIG. 6 shows a detailed operation example in the electrical angularcycle T0 of FIG. 5 and shows an operation example in the case wheremagnetization variation occurs as in the case of FIG. 25. In FIG. 6,similarly to FIG. 25, the detection of the phase error PERR is performedwith the zero crossing timing of the rising edge of the three-phase BEMFoccurring three times in the electrical angular cycle and the zerocrossing timing of the falling edge occurring three times in theelectrical angular cycle as the detection timings. At each detectiontiming, magnetization variation occurs as in the case of FIG. 25 but,unlike the case of FIG. 25, the phase error PERR is zero.

FIG. 7 is a block diagram showing a configuration example in which thePLL control loop of FIG. 3 is modified. The configuration example ofFIG. 7 differs from the configuration example of FIG. 3 in theconfiguration of the PI compensator PICPb. The PI compensator PICPbcomprises a phase error register unit REGPU instead of one phase errorregister REGP in the PI compensator PICPa of FIG. 3. The phase errorregister unit REGPU includes 24 phase error registers REGP0_0 to REGP3_5corresponding to a total of 24 detection timings in the mechanicalangular cycle, similarly to the cycle setting register unit REGNUa. Thestorage of the phase error PERR in the phase error registers REGP0_0 toREGP3_5 is controlled by the write enable signal WEN in the same manneras in the cycle setting register of FIG. 3.

The phase error register unit REGPU includes a selector SELP. As in theselector SELN of FIG. 3, the selector SELP selects a corresponding oneof the plurality of phase error registers REGP0_0 to REGP3_5 at eachdetection timing, and outputs the phase error PERR stored in theselected phase error register.

As a result, the PI compensator PICPb stores the phase error PERR, whichis the other one of the state variables, in the corresponding one of thephase error registers at each of the plurality of detection timings inthe previous mechanical angular cycle. Then, the PI compensator PICPbreflects the phase error PERR stored in the corresponding one of thephase error registers for each of the plurality of detection timings inthe current mechanical angular cycle and determines the cycle settingvalue (the amount of operation) NCNT in the current mechanical angularcycle.

As described above, although the PI compensator can be realized invarious configurations, in this embodiment, the PI compensator isconfigured to determine (calculate) the present cycle setting value NCNT(NCNTS by reflecting the previous phase error PERR) and the previouscycle setting value NCNT (NCNTS). Therefore, in order to perform the PIcontrol strictly, as in the case of the previous phase error PERR, thephase error PERR at the corresponding detection timing in the previousmechanical angular cycle needs to be used instead of the detectiontiming before the phase switching interval (electrical angle of 60°) asin the case of the cycle setting value NCNT(NCNTS).

Therefore, in FIG. 7, a phase error register unit REGPU is provided.When the method of the first embodiment is used, as shown in FIG. 6, itis considered that the phase error PERR at the respective detectiontimings are both close to zero. Therefore, even in the configurationexample of FIG. 3, a practical problem does not occur in particular. Byusing the configuration example of FIG. 3, the area overhead can bereduced as compared with the configuration example of FIG. 7. However,for example, in an application in which load fluctuation or the like isfrequent, if one phase error register REGP is used as shown in FIG. 3,the phase error caused by load fluctuation and the phase error caused bymagnetization fluctuation is confused and stored in the phase errorregister REGP. Therefore, when the configuration example of FIG. 7 isused, the component of the phase error due to the magnetizationvariation can be eliminated from the stored value of each phase errorregister, and therefore, it is considered that a more stable controloperation can be performed as compared with the configuration example ofFIG. 3.

As described above, when the method of the first embodiment is used,even when the magnetization variation occurs, the phase error PERRinputted to the PLL control unit (PI compensator) can be made close tozero. In this state, the three-phase motor SPM is driven in a stateclose to the ideal state shown in FIG. 19, that is, a state in which thephase of the BEMF and the conduction timings are always substantiallysynchronized with each other. As a result, even when magnetizationvariation occurs, it is possible to reduce rotational jitter.

Further, as shown in FIGS. 3 and 4, by allowing the switching mode andthe fixed mode to be selected by the detection mode selection signalCNTMD, it is possible to cope with high-speed responses such as when themotor is started. That is, for example, it is possible to perform anefficient operation such as selecting a fixed mode during a period inwhich a high-speed response is important instead of a rotational jitterand thereafter selecting a switching mode during a period of asteady-state rotation in which a reduction of the rotational jitter isimportant, such as at the time of starting a motor.

Here, in the fixed mode, the same cycle setting value NCNT is stored inall the cycle setting registers REGN0_0 to REGN3_5 by the control of thewrite enable signal WEN as described with reference to FIG. 4. As aresult, since the initial values of the cycle setting registers REGN0_0to REGN3_5 are set to appropriate values to some extent at the time ofsubsequently shifting to the switching mode, it is possible to smoothlyshift to the operation of the switching mode.

Here, in the 8-pole motor, the phase error PERR is detected at a totalof 24 detection timings included in the mechanical angular cycle. Sincethe magnetization variation is caused by mechanical factors, it has aperiodicity of a mechanical angle of 360°. In addition, at eachdetection timing within the mechanical angular cycle, the occurrencestate of magnetization variation may differ individually. Therefore, itis desired to detect the phase error PERR at a total of 24 detectiontimings. However, in this case, it is necessary to provide a total of 24high impedance periods (i.e. non-conduction periods) within themechanical angular cycle. Since the non-conduction period causesdisturbance of the driving current waveform, the non-conduction periodmay be desired to be reduced.

In such cases, for example, the falling zero crossing of the u phase,the v phase and the w phase may be excluded and the phase error PERR maybe detected (i.e. the phase error PERR may be detected at an interval of120° electrical angular) with the rising zero crossing as an object. Inthis case, for example, the write enable signal WEN corresponding toevery other cycle setting register REGN0_1, 0_3, 0_5, . . . may be fixedto the negate level. In addition, the PLL timing control unit PLLTC ofFIG. 24 may be configured to control the high-impedance signal HIZ andthe masking signal MSK based on the OR operation result of the writeenable signals WEN (HIZ=‘L’ and MSK=‘H’ when the OR operation result is‘L’ level).

Also, in some cases, for example, it is possible to target only therising zero crossing of the u phase and not the others, or to targetonly the falling zero crossing of the u phase and not the others. Forexample, with respect to the former, in the mechanical angular cycle ofthe “2×N (N is an integer of 2 or more)” pole motor, the occurrencestate of magnetization variation may be different in each of the Nrising zero crossings of the u phase, which may occur every electricalangular cycle. Therefore, by applying the method of the firstembodiment, an effect of reducing rotational jitter can be obtained ascompared with the configuration example of FIG. 21.

In addition, in the switching mode, in a state where the steady-staterotation is maintained, the stored values of the cycle setting registersREGN 0_0 to REGN3_5 are substantially constant, ideally unchanged.Therefore, in the switching mode, the feedback control by the PLL doesnot necessarily have to be always in the enable state. Specifically, thePLL control loop can perform the operations as shown in FIGS. 5 and 6,for example, only during an initial period after the steady-staterotation is reached, or intermittently thereafter.

In this instance, the PLL control loop searches for an appropriate cyclesetting value NCNT for each of the cycle setting registers REGN0_0 toREN3_5 by an operation as shown in FIGS. 5 and 6, for example, in aninitial period or the like, and then fixes the write enable signal WENat an ‘L’ level, for example. In response to this, the PLL timingcontrol unit PLLTC fixes the high-impedance signal HIZ to the ‘L’ leveland the masking signal MSK to the ‘H’ level. In this condition, the PLLcontrol loop may generate the phase switching signal COMM by using thesearch values of the cycle setting registers REGN0_0 to REN3_5. Thus,the influence of the non-conduction period can be reduced.

Second Embodiment

FIG. 8 is a block diagram showing a schematic configuration example of amotor driving apparatus according to Embodiment 2 of the presentinvention. The motor driving apparatus shown in FIG. 8 is different fromthe motor driving apparatus shown in FIG. 2 in that the parametersetting register PREGb outputs the window selection signal WNMD and thatthe window selection signal WNMD is input to the output control unitOTCT, specifically, the PLL timing control unit PLLTC.

FIG. 9 is a circuit diagram showing a configuration example around thePLL timing control unit in FIG. 8. The PLL timing control unit PLLTCashown in FIG. 9 has selectors SELW1, SELW2 added to the PLL-timingcontrol unit shown in FIG. 24. The selectors SELW1, SELW2 function as avariable setting unit that variably sets the length of the assertionperiod (masking period Tmsk) of the masking signal MSK and the length ofthe assertion period (high impedance period) of the high impedancesignal HIZ in accordance with the window selection signal WNMD.

In this embodiment, the selectors SELW1, SELW2 set the determinationvalues of the comparators CMPm1, CMPm2 to “24” and “32”, respectively,when the window selection signal WNMD is at the “L” level (‘0’ level)(referred to as the normal window mode). In this case, the PLL timingcontrol unit PLLTCa performs the same operation as in the case of FIG.24. On the other hand, when the window selection signal WNMD is at the“H” level (‘1’ level) (referred to as the shortened window mode), theselectors SELW1, SELW2 set the determination values of the comparatorsCMPm1, CMPm2 to “26” and “30”, respectively. As a result, the length ofthe masking period Tmsk is ½, and the length of the high-impedanceperiod is about ½.

In PLL control loop, the BEMF zero crossing timing needs to be includedat least in the masking period Tmsk (and the high-impedance period) todetect the phase error PERR. When the variation in magnetization occurs,the phase of the zero crossing timing varies, and therefore, the lengthof the masking period Tmsk needs to be ensured to a large extent by theamount of the variation. On the other hand, when the switching modedescribed in the first embodiment is used, even when the magnetizationvariation occurs, the zero crossing timing can be maintained at thereference timing (center timing) within the masking period Tmsk. As aresult, the length of the masking period Tmsk and the high-impedanceperiod can be shortened.

FIG. 10 is a timing chart showing an example of the operation of themotor driving apparatus shown in FIGS. 8 and 9. FIG. 10 shows anexemplary operation in the shortened window mode, in which the length ofthe masking period Tmsk and the length of the high-impedance period areboth shortened as compared with the case of FIG. 6. Along with theshortening of the high impedance period, for example, the drivingterminal voltage Vu based on the SP2 pattern is generated in the u-phasedriving terminal in the period of the difference before and after theshortening. As described above, in FIG. 10, the non-conduction period isshortened as compared with the case of FIG. 6 and the conduction periodis expanded accordingly. As an actual method of using the windowselection signal WNMD, for example, until the three-phase motor SPMreaches the steady-state rotation and stabilizes to some extent thesteady-state rotation, the normal window mode may be used and theshortened window mode may be switched to the shortened window mode at astage where the three-phase motor SPM is stabilized to some extent.

As described above, by using the method of the second embodiment, inaddition to the various effects described in the first embodiment, thetorque ripple can be further reduced and consequently, the noise andvibration of the three-phase motor SPM can be reduced. That is, the highimpedance period (non-conduction period) causes an increase in torqueripple, which may be a cause of noise or vibration of the three-phasemotor SPM, but by shortening the non-conduction period, such a problemcan be alleviated.

Third Embodiment

FIG. 11 is a circuit diagram showing a configuration example around thePLL timing control unit in the motor driving apparatus according to thethird embodiment of the present invention. In the PLL timing controlunit PLLTCb shown in FIG. 11, a selector SELH is added to theconfiguration shown in FIG. 9 and the AND gate ADh1 a for generating thehigh-impedance signal HIZ is changed from 2 inputs to 3 inputs. Theselector SELH outputs signals to one input to which the AND gate ADhlais added. The selector SELH and the AND gate ADhla function as ahigh-impedance releasing unit that releases the high-impedance statewhen the timing of the rising zero crossing or the timing of the fallingzero crossing of the BEMF occurs.

In this embodiment, the selector SELH outputs an “H” level which is afixed level when the window selection signal WNMD is at an “L” level(normal window mode). In this case, the same operation as that in FIG.24 is performed. On the other hand, when the window selection signalWNMD is at the “H” level (shortened window mode), the selector SELHselects and outputs the zero crossing EOR signal ZCEOR from the phaseerror detection unit PHED in the PLL control loop. As a result, the ANDgate ADhla negates the high-impedance signal HIZ to the “L” level(releases the high-impedance condition) at the time when the zerocrossing EOR signal ZCEOR is asserted to the “L” level (i.e. at the timewhen the zero crossing timing occurs).

FIG. 12 is a timing chart showing an example of the operation of themotor driving apparatus provided with the PLL timing control unit ofFIG. 9. In FIG. 12, the length of the masking period Tmsk is the same asthat in FIG. 10, but the length of the high impedance period isshortened by the high impedance releasing unit described above. Alongwith the shortening of the high impedance period, in FIG. 12, thenon-conduction period is further shortened as compared with the case ofFIG. 10 and the conduction period (in this example, the applicationperiod of the u-phase driving terminal voltage Vu based on the SP2pattern) is expanded accordingly.

After the detection of the zero crossing timing within the maskingperiod Tmsk, in particular, the BEMF needs not be monitored but only thecounting down operation of the error counters ECUNT is performed by thephase error detection unit PHED. Therefore, if the masking period Tmskis maintained, the high-impedance state is released in accordance withthe zero crossing timing and the conduction is restarted without anyparticular problems.

As described above, by using the method of the third embodiment, inaddition to the various effects described in the second embodiment, thetorque ripple can be further reduced and as a result, the three-phasemotor SPM can be further reduced in noise and vibration. Note that themethod of the third embodiment can be used in combination with theoperation example of FIG. 6 or the operation example of FIG. 25 in somecases. However, when combined with the operation example of FIG. 25,since the zero crossing timing fluctuates with respect to the referencetiming within the masking period Tmsk due to the influence of themagnetization variation, the effect of torque ripple reduction maydiffer for each phase switching period. In this respect, it is desirableto apply the present invention to the operation example of FIG. 10 orthe operation example of FIG. 6.

Fourth Embodiment

FIG. 13 is a block diagram showing a detailed configuration examplearound the PLL control loop of FIG. 2 in the motor driving apparatusaccording to the fourth embodiment of the present invention. Theconfiguration example shown in FIG. 13 differs from the configurationexample shown in FIG. 3 in the configuration of the cycle settingregister unit REGNUc in the PI compensator PICPc. In the PI compensatorPICPc, the PI calculation unit PICAL corresponds to the PI calculationunits PICALa, PICALb shown in FIGS. 3 and 7.

In the PI compensator PICPc, unlike the configuration of FIG. 3, one ofthe cycle setting registers REGN0_0 to REGN3_5 (here, REGN0_0) isdefined as a reference register. Furthermore, the PI compensator PICPccomprises a subtractor SBn and an adder ADDn. The subtractor SBncalculates a difference value between a state variable (here, the cyclesetting value NCNT) updated at each of a plurality of detection timingsand a state variable (cycle setting value NCNT) stored in the referenceregister.

When the reference register is selected based on the write enable signalWEN, the reference register REGN0_0 stores the cycle setting value NCNTfrom the PI calculation unit PICAL, similarly to the configurationexample of FIG. 3. On the other hand, when one cycle setting register(REGN0_1 to REGN3_5) excluding the reference register is selected basedon the write enable signal WEN, the subtractor SBn stores the calculateddifference value in the selected one cycle setting register.

When one register excluding the reference register is selected by theselector SELN, the adder ADDn adds the difference value from the oneregister and the state variable (cycle setting value NCNT) stored in thereference register and outputs the addition result as the cycle settingvalue NCNTS.

On the other hand, when the reference register is selected by theselector SELN, the selector SELN outputs “0” as the fixed input valueand in response to this, the adder ADDn outputs the state variable(cycle setting value NCNT) stored in the reference register as the cyclesetting value NCNTS as it is.

By using the method of the fourth embodiment, in addition to the variouseffects described in the first embodiment, the bit widths of the cyclesetting registers (REGN0_1 to REGN3_5) except for the reference register(REGN0_0) are narrowed along with the storage of the difference valuesso that the cycle setting register unit REGNUc can be reduced in areaand costs. The method of the fourth embodiment can also be applied tothe phase error register unit REGPU of FIG. 7.

Fifth Embodiment

FIG. 14 is a block diagram showing a schematic configuration example ofa motor driving apparatus according to Embodiment 5 of the presentinvention. In the above-described embodiments, examples of applicationto a position sensor-less motor for detecting BEMF have been described,but the method of the embodiment is not limited to a positionsensor-less motor and can be applied to a motor with a position sensor.FIG. 14 shows, as an example, a motor driving apparatus for athree-phase motor SPM which outputs three-phase Hall signals Hu, Hv andHw as three-phase position detection signals by a Hall element servingas a position sensor. The three-phase position detecting signal from theposition sensor is not limited to a Hall signal and may be varioussignals such as a UVW signal from an Resolver Digital Converter.

The motor driving apparatus shown in FIG. 14 does not have the BFMFdetection unit BFDET and the PWM fixing unit PWMFC because the positionis detected using the Hall signals Hu, Hv and Hw, unlike theconfiguration shown in FIG. 2. As a result, the configurations of thephase error detection unit PHEDd and the output control unit OTCTd areslightly different from those of the configuration shown in FIG. 2 andthe BEMF threshold value Vthb and the zero crossing mode signal ZCMD aredeleted from the output of the parameter setting register PREGd. Thephase error detection unit PHEDd detects the phase error PERR using theHall signals Hu, Hv and Hw instead of the zero crossing signal ZCOUT inFIG. 2. Unlike the configuration of FIG. 2, the output control unitOTCTd does not output the BEMF sampling signal BSPL or the phaseselection signal SEL, nor does it perform an operation corresponding tothe PWM-fixed signal PWMFIX.

FIG. 15 is a block diagram showing a detailed configuration examplearound the PLL control loop in FIG. 14. FIG. 16 is a timing chartshowing an example of the operation of the motor driving apparatus shownin FIGS. 14 and 15. In FIG. 15, the configuration of the phase errordetection unit PHEDd is different from that in FIG. 3. The phase errordetection unit PHEDd includes a zero crossing signal generation unit ZCGthat receives the Hall signals Hu, Hv, Hw and the BEMF polarity signalDIR and generates a zero crossing EOR signal ZCEOR similar to that inFIG. 3 and an error counter ECUNT similar to that in FIG. 3.

As shown in FIG. 16, the Hall signals Hu, Hv and Hw are, ideally,rectangular wave signals each having a rising edge and a falling edgesynchronized with the zero crossing timing of the three-phase backelectromotive force Vbemf (U, V, W). The phase error detection unitPHEDd performs a logical operation on the Hall signals Hu, Hv and Hw andthe BEMF polarity signal DIR to generate the zero crossing EOR signalZCEOR similar to the zero crossing EOR signal in FIG. 6. Thus, the errorcounter ECUNT can detect the phase error PERR in the same manner as inFIG. 6. Although the phase position signal CNT6 is generated using thephase position counter CUNTph in FIG. 15, the phase position signal CNT6can be generated by latching the decoding results of the Hall signalsHu, Hv and Hw with the phase switching signal COMM.

Here, the Hall signals Hu, Hv and Hw are ideally signals synchronizedwith the phase of the back electromotive force Vbemf (U, V, W). However,in the actual Hall signals Hu, Hv and Hw, as shown in FIG. 16, variation(i.e. magnetization variation) may occur with respect to the BEMF phaseof the corresponding phase. For example, the phase of the v-phase Hallsignals Hv is advanced of the phase of the v-phase back electromotiveforce Vbemf (V). Such variation in magnetization is caused by, forexample, variation in mounting of the Hall element. The variation inmagnetization causes rotational jitter in the same manner as in thefirst embodiment and the like. Therefore, as shown in FIG. 15, it isadvantageous to use the same PI compensator PICPa as in FIG. 3 and thelike.

As described above, by using the method of Embodiment 5, it is possibleto obtain the various effects described in Embodiment 1 and the like inmotor systems of various types and applications in which variation inmagnetization can occur. In addition, if the rotational jitter can bereduced regardless of the magnetization variation, it is possible tomanufacture various motors at low cost and consequently, it is possibleto reduce the cost of the motor system and the like. The motor system isnot limited to the HDD but includes a DVD playback/recording device, aBlu-ray playback/recording device and the like. Further, in some cases,it is also possible to apply the method of the embodiment as a techniquefor performing higher-precision control in an inverter system of an airconditioner or the like.

Although the invention made by the present inventor has beenspecifically described based on the embodiment, the present invention isnot limited to the embodiment described above, and various modificationscan be made without departing from the gist thereof. For example, theabove-described embodiments have been described in detail in order toeasily understand the present invention and are not necessarily limitedto those having all the configurations described. In addition, a part ofthe configuration of one embodiment can be replaced with theconfiguration of another embodiment, and the configuration of anotherembodiment can be added to the configuration of one embodiment. It isalso possible to add, delete, or replace some of the configurations ofthe respective embodiments.

1. A motor driving apparatus for controlling a targeted rotational phasefor control so that a detected rotational phase obtained by detecting arotational phase of a three-phase motor and the targeted rotationalphase for control to determine a conduction timing of the three-phasemotor are synchronized, the motor driving apparatus having a phase errordetecting unit for detecting a plurality of phase errors between thedetected rotational phase and the targeted rotational phase for controlat each of a plurality of detection timings in a mechanical angularcycle that is a cycle of a mechanical angle of 360 degrees in thethree-phase motor, a compensator for determining an amount of operationfor bringing the plurality of phase errors closer to zero by taking theplurality of phase errors from the phase error detecting unit as aninput and by reflecting state variables in the previous mechanicalangular cycle so as to update the state variables, and an conductiontiming generating unit for sequentially controlling the targetedrotational phase for control based on the amount of operation from thecompensator, wherein the compensator determines the amount of operationcorresponding to each of the plurality of detection timings in theprevious mechanical angular cycle.
 2. The motor driving apparatusaccording to claim 1, wherein the compensator has a plurality ofregisters corresponding to each of the plurality of detection timings,stores each of the state variables in corresponding each of theplurality of registers at each of the plurality of detection timings inthe previous mechanical angular cycle and determines the amount ofoperation in the current mechanical angular cycle by reflecting thestate variable stored in the corresponding each of the plurality ofregisters at each of the plurality of detection timings in the currentmechanical angular cycle.
 3. The motor driving apparatus according toclaim 2, wherein the detected rotation phase is a phase of a backelectromotive force of the three-phase motor, and wherein each of theplurality of detection timings is a timing of a rising zero crossing inwhich the back electromotive force crosses the middle of the amplitudeof the back electromotive force the toward a high potential side, oreach of the plurality of detection timings is a timing of a falling zerocrossing in which the back electromotive force crosses the middle of theamplitude of the back electromotive force the toward a low potentialside.
 4. The motor driving apparatus of claim 3, wherein the pluralityof detection timings include a timing of the rising zero crossingsoccurring three times in an electrical angular cycle having anelectrical angle of 360 degrees and a timing of the falling zerocrossings occurring three times in the electrical angular cycle.
 5. Themotor driving apparatus according to claim 3, wherein the three-phasemotor is a “2×N” (N is an integer equal to or greater than 2) polemotor, the mechanical angular cycle includes N times an electricalangular cycle of 360 degrees, and the plurality of detection timingsinclude at least one of a timing of the rising zero crossing occurringin each of the electrical angular cycles of the back electromotive forceof any one phase in three phases, or a timing of the falling zerocrossing occurring in each of the electrical angular cycles.
 6. Themotor driving apparatus according to claim 3, further comprising: atiming control unit that determines a plurality of reference timingssynchronized with the targeted rotational phase for control atpredetermined phase intervals within the mechanical angular cycle;controls a driving input terminal of the three-phase motor correspondingto the phase to be detected of the back electromotive force to a highimpedance state at a predetermined phase interval including theplurality of the reference timings; and the phase error detection unitdetects a phase error between the reference timing and the timing of therising zero crossing occurring at the driving input terminal in the highimpedance state or a phase error between the reference timing and thetiming of the falling zero crossing occurring at the driving inputterminal in the high impedance state.
 7. The motor driving apparatusaccording to claim 6, wherein the timing control unit has a variablesetting unit for variably setting the length of the predetermined phaseinterval.
 8. The motor driving apparatus according to claim 6, whereinthe timing control unit includes a high impedance releasing unit thatreleases the high impedance state when the timing of the rising zerocrossing or the timing of the falling zero crossing occurs.
 9. The motordriving apparatus according to claim 3, wherein the compensator operatesin a first operation mode or a second operation mode, wherein in thefirst operation mode, the amount of operation is determined by using thecorresponding one register for each of the plurality of detectiontimings, and in the second operation mode, the state variable for eachof the plurality of detection timings is commonly stored in theplurality of registers, whereby the plurality of registers are regardedas one register to determine the amount of operation.
 10. The motordriving apparatus of claim 2, wherein the state variable is the amountof operation.
 11. The motor driving apparatus according to claim 2,wherein the state variable is the amount of operation and the phaseerror, and the compensator has a plurality of first registers serving asthe plurality of registers for storing the amount of operation and aplurality of second registers serving as the plurality of registers forstoring the phase error.
 12. The motor driving apparatus according toclaim 2, wherein one of the plurality of registers is a referenceregister, and the compensator further comprises: a subtractor forcalculating a difference value between the state variable updated ateach of the plurality of detection timings and the state variable storedin the reference register, and for storing the difference value in thecorresponding one register excluding the reference register; and anadder for adding the difference value from the corresponding oneregister excluding the reference register and the state variable storedin the reference register.
 13. The motor driving apparatus according toclaim 2, wherein the detected rotational phase is a phase of athree-phase position detection signal from a position sensor attached tothe three-phase motor, and each of the plurality of detection timings isa timing of a rising edge or a timing of a falling edge of thethree-phase position detection signal.
 14. A motor system comprising: adisk for storing data; a three-phase motor for rotating the disk; amotor driving apparatus for controlling a targeted rotational phase forcontrol so that a detected rotational phase obtained by detecting arotational phase of a three-phase motor and the targeted rotationalphase for control to determine a conduction timing of the three-phasemotor are synchronized wherein the motor driving apparatus has: a phaseerror detecting unit for detecting a plurality of phase errors betweenthe detected rotational phase and the targeted rotational phase forcontrol at each of a plurality of detection timings in a mechanicalangular cycle that is a cycle of a mechanical angle of 360 degrees inthe three-phase motor, a compensator for determining an amount ofoperation for bringing the plurality of phase errors closer to zero bytaking the plurality of phase errors from the phase error detecting unitas an input and by reflecting state variables in the previous mechanicalangular cycle so as to update the state variables, and a conductiontiming generating unit for sequentially controlling the targetedrotational phase for control based on the amount of operation from thecompensator, wherein the compensator determines the amount of operationcorresponding to each of the plurality of detection timings in theprevious mechanical angular cycle.
 15. The motor system of claim 14,wherein the compensator has a plurality of registers corresponding toeach of the plurality of detection timings, stores each of the statevariables in corresponding each of the plurality of registers at each ofthe plurality of detection timings in the previous mechanical angularcycle and determines the amount of operation in the current mechanicalangular cycle by reflecting the state variable stored in thecorresponding each of the plurality of registers at each of theplurality of detection timings in the current mechanical angular cycle.16. The motor system of claim 15, wherein the detected rotation phase isa phase of a back electromotive force of the three-phase motor, andwherein each of the plurality of detection timings is a timing of arising zero crossing in which the back electromotive force crosses themiddle of the amplitude of the back electromotive force the toward ahigh potential side, or each of the plurality of detection timings is atiming of a falling zero crossing in which the back electromotive forcecrosses the middle of the amplitude of the back electromotive force thetoward a low potential side.
 17. The motor system of claim 16, whereinthe three-phase motor is a “2×N” (N is an integer equal to or greaterthan 2)” pole motor, the mechanical angular cycle includes N times anelectrical angular cycle of 360 degrees, and the plurality of detectiontimings include at least one of a timing of the rising zero crossingoccurring in each of the electrical angular cycles of the backelectromotive force of any one phase in three phases, or a timing of thefalling zero crossing occurring in each of the electrical angularcycles.
 18. The motor system according to claim 16, a timing controlunit that determines a plurality of reference timings synchronized withthe targeted rotational phase for control at predetermined phaseintervals within the mechanical angular cycle; controls a driving inputterminal of the three-phase motor corresponding to the phase to bedetected of the back electromotive force to a high impedance state at apredetermined phase interval including the plurality of the referencetimings; and the phase error detection unit detects a phase errorbetween the reference timing and the timing of the rising zero crossingoccurring at the driving input terminal in the high impedance state or aphase error between the reference timing and the timing of the fallingzero crossing occurring at the driving input terminal in the highimpedance state.
 19. The motor system of claim 16, wherein thecompensator operates in a first operation mode or a second operationmode, wherein in the first operation mode, the amount of operation isdetermined by using the corresponding one register for each of theplurality of detection timings, and in the second operation mode, thestate variable for each of the plurality of detection timings iscommonly stored in the plurality of registers, whereby the plurality ofregisters are regarded as one register to determine the amount ofoperation.
 20. The motor system of claim 15, wherein the state variableis the amount of operation and the phase error, and the compensator hasa plurality of first registers serving as the plurality of registers forstoring the amount of operation and a plurality of second registersserving as the plurality of registers for storing the phase error.